1. Field of the Invention
The present invention relates to fiber optic rotation rate sensors. More particularly, this invention pertains to a fiber optic Sagnac interferometer arrangement for avoiding lock-in and scale factor nonlinearities at low rotation rates.
2. Description of the Prior Art
Printed publication DE-A1-3,144,162 describes a method for measuring nonreciprocal phase shifts in a fiber optic Sagnac interferometer ("ring interferometer") by readjusting the nonreciprccal phase shifts (e.g. on the basis of rates of rotation) by applying, on the one hand, a phase deviation having a specific amplitude that can be reversed with the frequency f.sub.0 =1/2t.sub.0 and, on the other hand, a saw-toothed voltage that likewise shifts the phase deviation, t.sub.0 designating the transit time of each of the light beams through the fiber coil, to the phase modulator located in the region of the fiber coil input. The gradient of the compensating or resetting saw-toothed voltage (phase ramp) corresponds to rotation rate and is, thus, proportional to .phi..sub.0 /t.sub.0 where .phi..sub.0 is the nonreciprocal phase shift caused, for example, by a rotational movement. In practice, however, it is barely possible to employ the so-called phase ramp resetting method described in the above-referenced publication. This is due to the fact that inertial rotation rate measurements require accuracies with respect to the reversible phase deviations, amplitude or gradient of resetting signal and proportionality or scale factor that analog circuit design cannot necessarily guarantee.
U.S. Pat. No. 4,705,399 teaches an advance in the above-mentioned ramp resetting method for fiber optic ring interferometers that operates entirely digitally for signal evaluation and conditioning. As a result, the required signal precision, particularly with respect to accuracy of the reversible phase deviations and ramp resetting signal, can be obtained by a unique allocation of an automatic measuring range reversal and a correction of the scale factors.
The functional principle of the digital phase ramp resetting method of the prior art is first explained with reference to FIG. 14. (FIG. 14 corresponds, in a simplified way, to the prior art disclosed in FIG. 14 of U.S. Pat. No. 4,705,399. cf. H. C. Lefevre et al. in Integrated Optics: "A practical solution for the Fiber-Optic Gyroscope", SPIE Vol. 719, Fiber Optic gyros, 1986.)
The schematic representation of FIG. 14 shows a light source L (e.g., a laser) whose parallel light beams are split into two light beams via a beam splitter ST1, and irradiated into an interferometer fiber coil FS in opposite directions. The fiber coil FS preferably consists of an optical monomode fiber. The beam splitter ST1 simultaneously acts as a mixer for recombining the two light beams after traverse of the fiber coil FS. The interference signal of the two superimposed light beams passes via a second beam splitter ST2 and the output branch AUS to a photodetector PD that scans the intensity of the interference pattern. Using .DELTA..phi..sub.0 to designate the phase difference between the two light beams that propagate in opposite directions in the closed fiber coil FS, it holds that .DELTA..phi..sub.0 =0 as long as there are no nonreciprocal disturbances. (Reference should be made to the cited literature concerning the mathematical relation between the phase difference .DELTA..phi..sub.0 and the rotation rate or rotational speed, the optical power density at the input of the photodetector PD, and measuring sensitivity.) A method for increasing the sensitivity of the interferometer by introducing a constant, nonreciprocal bias to the two counterrotating light beams in the fiber coil FS so that the counter-rotating beams (light waves) are periodically shifted (by a phase modulator PM) to the operating point of highest sensitivity of the interferometer by (2n+1).pi./2 where n is a whole number is also described. The phase modulator PM is first excited with a signal .phi..sub.1 (t) that causes a period phase shift (e.g. .+-..pi./2, 3/2.pi., . . . ,) of period 2t.sub.0, t.sub.0 designating the transit time of a light wave in the fiber coil FS.
In the circuit of FIG. 14, nonreciprocal phase shifts that result from negative feedback to the phase modulator PM are compensated, as described in the referenced patent, by a so-called phase ramp signal, whose gradient is proportional to .DELTA..phi./t.sub.0, where .DELTA..phi. is the nonreciprocal phase shift. The resetting phase ramp signal is a sawtooth or staircase waveform in which the sawtooth amplitude or riser height is equal to .DELTA..phi. and the duration of the sawtooth or riser corresponds to the transit time t.sub.0 or an odd-numbered multiple thereof.
In contrast to an analog solution the digital phase ramp principle described, e.g., in U.S. Pat. No. 4,705,399, possesses a decided advantage in that the scale factors of the phase modulation control and phase ramp resetting signals are corrected, at least in principle, and precise definition of the sawtooth amplitude of the resetting signal is guaranteed.
The function of a circuit in accordance with FIG. 14 (i.e., generating a reciprocal phase shift that alternates in time with the frequency f.sub.0 and optimizing the operating point) and the stair-step ramp resetting signal including scale factor regulation, is explained briefly as follows. The output VD of the photodetector PD is raised by an impedance converter and amplifier A.sub.0, whose output signal VD' feeds a synchronous demodulator SYNCD. The demodulator is synchronized to the scanning frequency f.sub.0 =1/2t.sub.0. Its output passes as the signal VA through an amplifier A that, as a rule, is combined with a filter, to an analog-to-digital converter AD. The digital output of the converter is proportional to rotation rate and contains an item of sign information with respect to the direction of rotation. The signal VAD then passes to a circuit GSC, that essentially comprises a digital integrator that supplies a combined phase control signal VSC that consists of the digital phase ramp resetting signal and the digital phase modulation signal. The combined digital signal VSC is subsequently converted in a digital-to-analog converter DA to an analog control voltage VC, and transmitted by a driver amplifier AP as control voltage to the phase modulator PM.
The GSC circuit comprises a first memory M for intermediate storage of the digital signal VAD. The intermediate stored output signals SM.sub.1 pass to a first input e.sub.1 of an adder ADD that is connected in cascade to a second memory M.sub.2. Output signals VSC stored intermediately in the second memory M.sub.2 are fed back to a second input e.sub.2 of the adder ADD and added to the digital value of the rotation rate signal. The output signal SADD of the adder ADD thus corresponds to the angle of rotation.
The circuit described so far is synchronized and controlled by a central processor CPU through a bidirectional bus BC that is connected to a quartz-stabilized oscillator OSC that supplies the frequency f.sub.0.
Aside from the number of bits per word (referring to a specific angle of rotation, e.g. four seconds of arc, which bits correspond to the capacity of the adder ADD), the adder supplies an overflow signal at an output SL via the bus BC to the central processor CPU. In accordance with the overflow signals from the adder and the clock signal of the oscillator OSC the central processor CPU generates the various control and synchronization signals. A switchover is made between a "mode A" and a "mode B" in order to enable a scale factor correction, that depends upon the overflow of the modulation deviation of adder ADD. The switchover is made in such a way that a modulation deviation of, for example, .+-..pi./2 holds for mode A and .+-.3.pi./2 holds for mode B. As is described in the above-referenced literature, detected amplitude differences between the operating states with different modulation deviations serve as a measure of the scale factor error at phase ramp signal overflow. To correct such scale factor error, the circuit of FIG. 14 is equipped with a demodulator SFC that detects amplitude differences between the f.sub.0 modulation signal in mode A (phase deviation e.g. .+-..pi./2) and mode B (phase deviation e.g. .+-.3.pi./2). The SFC transmits the demodulated signal as an analog correction signal SIA via an integrating amplifier IA to a correcting analog input e.sub.m of the digital-to-analog converter DA.
The functional principle of the digital ramp resetting method for fiber optic rate-of-rotation sensors in which the gradient of the phase ramp is a measure of the reset rotation rate and in which there is a reversal of the modulation deviation in a ratio 1:3 in the return phase, in order to obtain a correction signal for the scale factor, is described briefly with reference to FIG. 14 and leads, in practical operation of such parasitic rotation rate measuring devices to substantial difficulties as discussed below.
The reversal from "mode A" to "mode B" is directly dependent upon the ramp value of the resetting sawtoothed signal and, thus, the angle of rotation of the gyroscope. In this regard, a ramp traversal can correspond to an angular increment of, for example, approximately four seconds of arc. However, by trebling the amplitude of the modulation signal and through parasitics in the sensitive signal path of the photodetector signal VD, modulation deviation reversal causes reversal of the gyroscope bias. Such undesired parasitic effects are indicated by dashed lines and the coupling factor K of FIG. 14. Since modulation deviation signal reversal is, however, dependent upon the angle of rotation, this leads to gyroscope deadband or lock-in. As explained below, this effect also leads to a scale factor nonlinearity outside the lock-in zone.
As is shown in FIG. 13 of U.S. Pat. No. 4,705,399, the mean intensity I of the photodetector signal VD will differ in the case of modulation deviations of mode A for that in the case of modulation deviations of mode B given the occurrence of a scale factor error. This intensity difference is integrated in the integrating amplifier IA and supplies the analog correction signal at the input e.sub.m of the digital-to-analog converter DA. The frequency of change in intensity is equal to the frequency of change from mode A to mode B, and thus equals the ramp signal return frequency as reversal (e.g., from mode A to mode B), is produced by ramp overflow (i.e., the signal SL of the adder ADD). This frequency of change is proportional to the rotation rate. That is, in accordance with the example explained in the referenced patent, a return of 2.pi. corresponds to an angular increment of 5 seconds of arc; at a 1 Hz return frequency this corresponds to a rotation rate of 5.degree./h.
In the case of low rotation rates, the frequency of change can become arbitrarily low. Accordingly, the integrating amplifier IA seldom contains intensity difference information. During the arbitrarily long intervals during which the mode does not change, the integrating amplifier IA does not possess such information. Thus, every small electrical zero error at its input causes its output quantity (the scale factor correction signal SIA) to drift. The scale factor is, therefore, susceptible to drifting of the integrating amplifier IA at low rotation rates. The percentage error in the absolute value of rotation rate caused by scale factor drift is small for low rotation rates. However, a sudden sharp rise in rotation rate becomes problematic since the scale factor is still "wrong" and, thus, large errors (absolute value) also will occur in the rotation rate at least until the scale factor control circuit returns to steady state.
It would, therefore seem advisable to remove, or at least reduce, the electromagnetic parasitics (disturbances with a coupling factor K; see FIG. 14) by electromagnetic compatibility measures (EMC measures). That is, one might consider the use of shields (indicated in FIG. 14 by shielding the connecting line from the driver amplifier AP to the phase modulator PM) and the installation of filters in the signal and voltage feed lines. However, the known interferometer design of FIG. 14 presents special EMC problems. The signal VSC or VC, VC' contains the modulation frequency f.sub.0 =1/2t.sub.0 that is generated in the oscillator OSC or via the processor CPU. However, the photodetector signal VD contains the rotation rate information with the same frequency and phase angle. This signal is detected by the synchronous demodulator SYNCD. The circuit groups that generate the modulation quantity of frequency f.sub.0, and the circuit parts that conduct the signal of the same frequency which is sensitive to rotation rate, are closely connected in space and have, by and large, to be fed from a common power supply device. This clearly presents a danger that electromagnetic energy of frequency f.sub. 0 will parasitically enter the sensitive signal path (signal VD). The use of stop filters for f.sub.0 in the signal lines is not possible since the desired signal information is present at such frequency. Undesired parasitics can be reduced only to a certain extent by shielding the amplifier A.sub.0 and the synchronous demodulator SYNCD against the rest of the circuit and by filtering their power supply.
A numerical example of the parasitic sensitivity will immediately clarify the particular problems to a person skilled in the art. The spectral component of frequency f.sub.0 in the signal VC or VC' is generally located in the range of a few V. By contrast, in the rotation rate signal VD voltages in the range of a few nV correspond--depending upon optical power, detector sensitivity and gyroscope scale factor--to a rotation rate of 1.degree./h. With such large amplitude differences, an undesired parasitic path will be unavoidable despite all possible EMC measured, as is symbolized in FIG. 12 by the coupling factor K located between the signals VC, VC' and VD. Naturally, due to different parasitic amplitudes, different errors, equivalent to rotation rates, in the signal VD, will be dependent upon the operating state (mode A or B) since, as assumed, the latter has a modulation amplitude three times higher than the former, and thus has a stronger parasitic effect. The different parasitic amplitudes lead, with reference to rotation rates, to different gyroscope zero point errors that are designated as biases B.sub.a or B.sub.b, depending upon the instantaneous modulation state (mode A or mode B).
Thus, the ramp gradient of the resetting signal is regulated by the closed control circuit proportional to the sum of the true input rotation rate D.sub.e and the respective bias B.sub.a or B.sub.b, depending upon modulation mode.
FIG. 1 shows an example in which D.sub.e +B.sub.a &gt;O, D.sub.e +B.sub.b &gt;O and B.sub.a &gt;B.sub.b (i.e., different biases B.sub.a or B.sub.b are present). Assuming that D.sub.e is a constant, different ramp gradients are produced depending upon whether mode A or mode B is present at that instant, since ramp gradient is proportional to D.sub.e +B.sub.a or D.sub.e +B.sub.b.
In the example of FIG. 1, the gradient during mode B is flatter than in mode A, since B.sub.b &lt;B.sub.a. Thus, the dwell time t.sub.b in mode B is larger than the dwell time t.sub.a in mode A than would be the case for biases independent of mode (B.sub.a =B.sub.b). However, since not only B.sub.a and B.sub.b but also the input rate of rotation D.sub.e influences the ramp gradient in both modes, the pulse-duty factor of the modes (T.sub.a /T.sub.b) also depends upon D.sub.e. The scale factor nonlinearity mentioned above results from this.
On the other hand, the above-mentioned lock-in effect occurs in a rotation rate range for which the sign of the sum (D.sub.e +B.sub.a) or (D.sub.e +B.sub.b) differs for the two modes (e.g., (D.sub.e +B.sub.a)&gt;O or (D.sub.e +B.sub.b &lt;O). This case, entirely realistic for low rotation rates, is illustrated in FIG. 2. The ramp of the resetting signal (illustrated without modulation for clarity) starts, for example, in mode A with a positive gradient, since (D.sub.e +B.sub.a)&gt;O. When the ramp reaches the (upper) overflow range (overflow signal SL in FIG. 14), a switch is made to mode B. However, the control then triggers a negative ramp gradient, since it now holds that (D.sub.e +B.sub.b)&lt;O. Thus, the ramp gradient changes its sign and leaves the overflow region and mode A once again holds. However, a positive ramp gradient is once again associated with mode A until a reversal to mode B once again takes place. Thus, the control is held "captive" and the interferometer is in the lock-in state. How quickly the " zig-zag-ramp" represented in FIG. 2 changes at the overflow limit between modes A and B is determined by the rapidity of the control which is generally very high. The lock-in state holds for an input rotation rate range in which the specified inequalities are fulfilled. The lock-in range thus has a width of .vertline.B.sub.a -B.sub.b .vertline..
In the description so far, the ramp value is the instantaneous value of the (unmodulated) ramp .phi.(t). This is proportional to the angle of rotation (and corresponds to the signal SADD). The ramp gradient d.phi./dt corresponds to the time derivative of the angle of rotation or rotation rate. The problem of the ramp resetting method of the prior art is that the ramp value (value of the angle of rotation) is the criterion for operating mode A or B and whether the different biases B.sub.a or B.sub.b are present and, since B.sub.a .noteq.B.sub.b, influences the ramp gradient itself. The scale factor error information is present, as mentioned, in the form of a signal of change in intensity, whose frequency is proportional to the rotation rate that can vary between 0 and approximately 100 kHz when the theoretical pulse-duty factor of the modulation deviations between mode A and mode B is 1:3. The apparent advantage of information-carrying changes in intensity occurring more often at high rates of rotation and that, consequently, better scale factor error information is available at high rotation rates, is opposed, however, by the fact that the associated signal processing device has to process a comparatively high frequency range. However, as mentioned, the known ramp resetting method in the region of zero rotation rate leads, on the other hand, to the absence of the scale factor error information and thus to drift problems.